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19-3244; Rev 0; 4/04
KIT ATION EVALU BLE AVAILA
TFT-LCD DC-DC Converters with Operational Amplifiers
Features
2.6V to 5.5V Input Supply Range 1.2MHz Current-Mode Step-Up Regulator Fast Transient Response to Pulsed Load High-Accuracy Output Voltage (1.5%) Built-In 14V, 2.4A, 0.16 N-Channel MOSFET High Efficiency (90%) Linear-Regulator Controllers for VGON and VGOFF High-Performance Operational Amplifiers 150mA Output Short-Circuit Current 13V/s Slew Rate 12MHz, -3dB Bandwidth Rail-to-Rail Inputs/Outputs Logic-Controlled, High-Voltage Switch with Adjustable Delay Timer-Delay Fault Latch for All Regulator Outputs Thermal-Overload Protection 0.6mA Quiescent Current
General Description
The MAX1516/MAX1517/MAX1518 include a high-performance step-up regulator, two linear-regulator controllers, and high-current operational amplifiers for active-matrix thin-film transistor (TFT) liquid-crystal displays (LCDs). Also included is a logic-controlled, high-voltage switch with adjustable delay. The step-up DC-DC converter provides the regulated supply voltage for the panel source driver ICs. The converter is a high-frequency (1.2MHz) current-mode regulator with an integrated 14V n-channel MOSFET that allows the use of ultra-small inductors and ceramic capacitors. It provides fast transient response to pulsed loads while achieving efficiencies over 85%. The gate-on and gate-off linear-regulator controllers provide regulated TFT gate-on and gate-off supplies using external charge pumps attached to the switching node. The MAX1518 includes five high-performance operational amplifiers, the MAX1517 includes three, and the MAX1516 includes one operational amplifier. These amplifiers are designed to drive the LCD backplane (VCOM) and/or the gamma-correction divider string. The devices feature high output current (150mA), fast slew rate (13V/s), wide bandwidth (12MHz), and rail-to-rail inputs and outputs. The MAX1516/MAX1517/MAX1518 are available in 32pin thin QFN packages with a maximum thickness of 0.8mm for ultra-thin LCD panels.
MAX1516/MAX1517/MAX1518
Minimal Operating Circuit
VCN VCP
VIN LX IN STEP-UP CONTROLLER PGND COMP AGND FB
VMAIN
Applications
Notebook Computer Displays LCD Monitor Panels Automotive Displays
VCP
MAX1518
DRVP GATE-ON CONTROLLER FBP VGON
Ordering Information
PART MAX1516ETJ MAX1517ETJ MAX1518ETJ TEMP RANGE -40C to +100C -40C to +100C -40C to +100C PIN-PACKAGE 32 Thin QFN 5mm x 5mm 32 Thin QFN 5mm x 5mm 32 Thin QFN 5mm x 5mm
SRC DEL COM SWITCH CONTROL CTL VCN DRN
DRVN GATE-OFF CONTROLLER SUP NEG1 FBN VGOFF
OUT1
OP1 REF REF NEG4
POS1 NEG2
OUT2
OP2
OP4
OUT4
POS2
POS4 NEG5
OUT3
OP3
OP5
OUT5
POS3 BGND
POS5
Pin Configurations appear at end of data sheet. ________________________________________________________________ Maxim Integrated Products 1
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
ABSOLUTE MAXIMUM RATINGS
IN, CTL to AGND ......................................................-0.3V to +6V COMP, FB, FBP, FBN, DEL, REF to AGND ....-0.3V to (VIN + 0.3V) PGND, BGND to AGND ......................................................0.3V LX to PGND ............................................................-0.3V to +14V SUP to AGND .........................................................-0.3V to +14V DRVP, SRC to AGND..............................................-0.3V to +30V POS_, NEG_, OUT_ to AGND ...................-0.3V to (VSUP + 0.3V) POS1 to NEG1, POS2 to NEG2, POS3 to NEG3, POS4 to NEG4, POS5 to NEG5 ...............................-6V to +6V DRVN to AGND ...................................(VIN - 30V) to (VIN + 0.3V) COM, DRN to AGND ................................-0.3V to (VSRC + 0.3V) DRN to COM............................................................-30V to +30V OUT_ Maximum Continuous Output Current....................75mA LX Switch Maximum Continuous RMS Output Current .........1.6A Continuous Power Dissipation (TA = +70C) 32-Pin Thin QFN (derate 21.2mW/C above +70C) ..1702mW Operating Temperature Range .........................-40C to +100C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25A, TA = 0C to +85C. Typical values are at TA = +25C, unless otherwise noted.)
PARAMETER IN Supply Range IN Undervoltage-Lockout Threshold SYMBOL VIN VUVLO VIN rising, typical hysteresis = 150mV VFB = VFBP = 1.4V, VFBN = 0, LX not switching IN Quiescent Current IIN VFB = 1.1V, VFBP = 1.4V, VFBN = 0, LX switching CONDITIONS MIN 2.6 2.3 2.5 0.6 6 55 -2A < IREF < 50A, VIN = 2.6V to 5.5V Temperature rising Hysteresis VMAIN fOSC TA = +25C to +85C TA = 0C to +85C VIN 1020 84 VFB No load VFB falling 0 < IMAIN < full load, transient only VIN = 2.6V to 5.5V VFB = 1.4V ICOMP = 5A FB to COMP -40 75 150 600 1.221 1.218 0.96 1200 87 1.233 1.233 1.00 -1.6 +0.04 0.15 +40 280 1.231 1.250 +160 15 13 1380 90 1.245 1.247 1.04 1.269 TYP MAX 5.5 2.7 0.8 mA 11 ms V C UNITS V V
Duration to Trigger Fault Condition REF Output Voltage Thermal Shutdown MAIN STEP-UP REGULATOR Output Voltage Range Operating Frequency Oscillator Maximum Duty Cycle FB Regulation Voltage FB Fault Trip Level FB Load Regulation FB Line Regulation FB Input Bias Current FB Transconductance FB Voltage Gain
V kHz % V V % %/ V nA S V/ V
2
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TFT-LCD DC-DC Converters with Operational Amplifiers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25A, TA = 0C to +85C. Typical values are at TA = +25C, unless otherwise noted.)
PARAMETER LX On-Resistance LX Leakage Current LX Current Limit Current-Sense Transconductance Soft-Start Period Soft-Start Step Size OPERATIONAL AMPLIFIERS SUP Supply Range SUP Supply Current VSUP MAX1518 ISUP Buffer configuration, VPOS_ = 4V, no load MAX1517 MAX1516 4.5 3.2 2 0.7 0 +1 0 0 (VNEG_, VPOS_) VSUP 45 125 IOUT_ = 100A Output Voltage Swing, High VOH IOUT_ = 5mA Output Voltage Swing, Low Short-Circuit Current Output Source and Sink Current Power-Supply Rejection Ratio Slew Rate -3dB Bandwidth Gain-Bandwidth Product FBP Regulation Voltage FBP Fault Trip Level FBP Input Bias Current FBP Effective Load-Regulation Error (Transconductance) IFBP GBW VFBP RL = 10k, CL = 10pF, buffer configuration Buffer configuration IDRVP = 100A VFBP falling VFBP = 1.4V VDRVP = 10V, IDRVP = 50A to 1mA 1.231 0.96 -50 -0.7 PSRR VOL IOUT_ = -100A IOUT_ = -5mA To VSUP / 2, source or sink (VNEG_ , VPOS_, VOUT_) VSUP / 2, |VOS| < 10mV DC, 6V VSUP 13V, (VNEG_, VPOS_) VSUP/2 50 40 60 13 12 8 1.250 1.00 1.269 1.04 +50 -1.5 VSUP 15 VSUP 150 VSUP 3 mV VSUP 80 2 70 150 15 150 mV mA mA dB V/s MHz MHz V V nA % 13.0 4.8 3 1.1 12 50 VSUP mV nA V dB dB mA V tSS SYMBOL RLX(ON) ILX ILIM VLX = 13V VFB = 1V, duty cycle = 65% 2.5 3.0 CONDITIONS MIN TYP 160 0.02 3.0 3.8 14 ILIM / 8 MAX 250 40 3.5 5 UNITS m A A S ms A
MAX1516/MAX1517/MAX1518
Input Offset Voltage Input Bias Current Input Common-Mode Range Common-Mode Rejection Ratio Open-Loop Gain
VOS IBIAS VCM CMRR
(VNEG_, VPOS_, VOUT_) VSUP / 2, TA = +25C (VNEG_ , VPOS_, VOUT_) VSUP / 2
GATE-ON LINEAR-REGULATOR CONTROLLER
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3
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25A, TA = 0C to +85C. Typical values are at TA = +25C, unless otherwise noted.)
PARAMETER FBP Line (IN) Regulation Error DRVP Sink Current DRVP Off-Leakage Current Soft-Start Period Soft-Start Step Size GATE-OFF LINEAR-REGUALTOR CONTROLLER FBN Regulation Voltage FBN Fault Trip Level FBN Input Bias Current FBN Effective Load-Regulation Error (Transconductance) FBN Line (IN) Regulation Error DRVN Source Current DRVN Off-Leakage Current Soft-Start Period Soft-Start Step Size POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES DEL Capacitor Charge Current DEL Turn-On Threshold DEL Discharge Switch OnResistance CTL Input Low Voltage CTL Input High Voltage CTL Input Leakage Current CTL-to-SRC Propagation Delay SRC Input Voltage Range SRC Input Current SRC to COM Switch OnResistance DRN to COM Switch OnResistance COM to PGND Switch OnResistance ISRC RSRC(ON) RDRN(ON) RCOM(ON) VDEL = 1.5V, CTL = IN VDEL = 1.5V, CTL = AGND VDEL = 1.5V, CTL = IN VDEL = 1.5V, CTL = AGND VDEL = 1.1V 350 50 15 6 35 1000 VTH(DEL) During UVLO, VIN = 2.2V VIN = 2.6V to 5.5V VIN = 2.6V to 5.5V CTL = AGND or IN 2 -1 100 28 100 30 12 70 1800 +1 During startup, VDEL = 1V 4 1.19 5 1.25 20 0.6 6 1.31 A V V V A ns V A tSS IDRVN IFBN VFBN IDRVN = 100A VFBN rising VFBN = 0 VDRVN = -10V, IDRVN = 50A to 1mA IDRVN = 0.1mA, 2.6V < VIN < 5.5V VFBN = 500mV, VDRVN = -10V VFBN = 0V, VDRVN = -25V 1 235 370 -50 11 +0.7 4 -0.01 14 VREF / 128 -10 250 420 265 470 +50 25 5 mV mV nA mV mV mA A ms V tSS IDRVP SYMBOL CONDITIONS IDRVP = 100A, 2.6V < VIN < 5.5V VFBP = 1.1V, VDRVP = 10V VFBP = 1.4V, VDRVP = 28V 1 MIN TYP 1.5 5 0.01 14 VREF / 128 10 MAX 5 UNITS mV mA A ms V
4
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TFT-LCD DC-DC Converters with Operational Amplifiers
ELECTRICAL CHARACTERISTICS
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25A, TA = -40C to +85C, unless otherwise noted.) (Note 1)
PARAMETER IN Supply Range IN Undervoltage-Lockout Threshold SYMBOL VIN VUVLO VIN rising, typical hysteresis = 150mV VFB = VFBP = 1.4V, VFBN = 0, LX not switching IN Quiescent Current IIN VFB = 1.1V, VFBP = 1.4V, VFBN = 0, LX switching -2A < IREF < 50A, VIN = 2.6V to 5.5V VMAIN fOSC VFB No load VIN = 2.6V to 5.5V VFB = 1.4V ICOMP = 5A RLX(ON) ILIM VSUP MAX1518 SUP Supply Current Input Offset Voltage Input Common-Mode Range ISUP VOS VCM IOUT_ = 100A Output Voltage Swing, High VOH IOUT_ = 5mA Output Voltage Swing, Low Short-Circuit Current Output Source and Sink Current VOL IOUT_ = -100A IOUT_ = -5mA To VSUP / 2 Source Sink 50 50 40 Buffer configuration, VPOS_ = 4V, no load MAX1517 MAX1516 0 VSUP 15 mV VSUP 150 15 150 mV mA mA VFB = 1V, duty cycle = 65% 2.5 4.5 -40 75 1.222 VIN 1020 1.212 CONDITIONS MIN 2.6 2.265 MAX 5.5 2.715 0.8 mA 11 1.269 13 1380 1.250 0.15 +40 300 250 3.5 13.0 4.8 3.0 1.1 12 VSUP mV V mA V V kHz V %/ V nA S m A V UNITS V V
MAX1516/MAX1517/MAX1518
REF Output Voltage MAIN STEP-UP REGULATOR Output Voltage Range Operating Frequency FB Regulation Voltage FB Line Regulation FB Input Bias Current FB Transconductance LX On-Resistance LX Current Limit OPERATIONAL AMPLIFIERS SUP Supply Range
(VNEG_, VPOS_, VOUT_) VSUP / 2
(VNEG_ , VPOS_, VOUT_) VSUP / 2, |VOS| < 10mV VFBP IDRVP = 100A
GATE-ON LINEAR-REGULATOR CONTROLLER FBP Regulation Voltage 1.218 1.269 V
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5
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25A, TA = -40C to +85C, unless otherwise noted.) (Note 1)
PARAMETER FBP Effective Load-Regulation Error (Transconductance) FBP Line (IN) Regulation Error DRVP Sink Current FBN Regulation Voltage FBN Effective Load-Regulation Error (Transconductance) FBN Line (IN) Regulation Error DRVN Source Current DEL Capacitor Charge Current DEL Turn-On Threshold CTL Input Low Voltage CTL Input High Voltage SRC Input Voltage Range SRC Input Current SRC to COM Switch OnResistance DRN to COM Switch OnResistance COM to PGND Switch OnResistance ISRC RSRC(ON) RDRN(ON) RCOM(ON) VDEL = 1.5V, CTL = IN VDEL = 1.5V, CTL = AGND VDEL = 1.5V, CTL = IN VDEL = 1.5V, CTL = AGND VDEL = 1.1V 350 VTH(DEL) VIN = 2.6V to 5.5V VIN = 2.6V to 5.5V 2 28 100 30 12 70 1800 IDRVN IDRVP VFBN SYMBOL CONDITIONS VDRVP = 10V, IDRVP = 50A to 1mA IDRVP = 100A, 2.6V < VIN < 5.5V VFBP = 1.1V, VDRVP = 10V IDRVN = 100A VDRVN = -10V, IDRVN = 50A to 1mA IDRVN = 0.1mA, 2.6V < VIN < 5.5V VFBN = 500mV, VDRVN = -10V During startup, VDEL = 1V 1 4 1.19 6 1.31 0.6 1 235 265 25 5 MIN MAX -2 5 UNITS % mV mA mV mV mV mA A V V V V A
GATE-OFF LINEAR-REGULATOR CONTROLLER
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
Note 1: Specifications to -40C are guaranteed by design, not production tested.
6
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TFT-LCD DC-DC Converters with Operational Amplifiers
Typical Operating Characteristics
(Circuit of Figure 1. VIN = 5V, VMAIN = 13V, VGON = 24V, VGOFF = -8V, VOUT1 = VOUT2 = VOUT3 = VOUT4 = VOUT5 = 6.5V, TA = +25C unless otherwise noted.)
STEP-UP EFFICIENCY vs. LOAD CURRENT
MAX1516 toc01
MAX1516/MAX1517/MAX1518
SWITCHING FREQUENCY vs. INPUT VOLTAGE
MAX1516 toc02
STEP-UP SUPPLY CURRENT vs. SUPPLY VOLTAGE
NO LOAD, SUP DISCONNECTED, R1 = 95.3k, R2 = 10.2k 8 SUPPLY CURRENT (mA) CURRENT INTO INDUCTOR
MAX1516 toc03
100 90 80 EFFICIENCY (%) 70 60 50 40 30 1 10 100 VOUT = 13V VIN = 5.0V VIN = 3.3V
1.4 SWITCHING FREQUENCY (MHz)
10
1.3
6
1.2
4
CURRENT INTO IN PIN
1.1
2
1.0 1000 2.5 3.0 3.5 4.0 4.5 5.0 5.5 LOAD CURRENT (mA) INPUT VOLTAGE (V)
0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 SUPPLY VOLTAGE (V)
STEP-UP REGULATOR SOFT-START (HEAVY LOAD)
MAX1516 toc04
STEP-UP REGULATOR PULSED LOAD-TRANSIENT RESPONSE
MAX1516 toc05
A 0V B 0V 13V B C 0A A 200mA
C 0A 2ms/div A: VIN, 5V/div B: VMAIN, 5V/div C: INDUCTOR CURRENT, 1A/div 10s/div A: LOAD CURRENT, 1A/div B: VMAIN, 200mV/div, AC-COUPLED C: INDUCTOR CURRENT, 1A/div
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TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
Typical Operating Characteristics (continued)
(Circuit of Figure 1. VIN = 5V, VMAIN = 13V, VGON = 24V, VGOFF = -8V, VOUT1 = VOUT2 = VOUT3 = VOUT4 = VOUT5 = 6.5V, TA = +25C unless otherwise noted.)
TIMER DELAY LATCH RESPONSE TO OVERLOAD
MAX1516 toc06
REF VOLTAGE LOAD REGULATION
MAX1516 toc07
GATE-ON REGULATOR LINE REGULATION
MAX1516 toc08
1.253 A 0V REF VOLTAGE (V) 1.251 1.250 1.249 1.248 0A 1.247 0 10 20 30 40 1.252
0.2 0 -0.2 -0.4 -0.6 -0.8 -1.0 VGON = 23.5V IGON = 20mA 23 24 25 26 27 28 29
B 55ms
10ms/div A: VMAIN, 5V/div B: INDUCTOR CURRENT, 1A/div
50
OUTPUT VOLTAGE ERROR (%)
30
LOAD CURRENT (A)
INPUT VOLTAGE (V)
GATE-ON REGULATOR LOAD REGULATION
MAX1516 toc09
GATE-OFF REGULATOR LINE REGULATION
MAX1516 toc10
GATE-OFF REGULATOR LOAD REGULATION
MAX1516 toc11
0 -0.05 VOLTAGE ERROR (%) -0.10 -0.15 -0.20 -0.25 -0.30 0 5 10 LOAD CURRENT (mA) 15
1.00 VGOFF = -8V IGOFF = 50mA OUTPUT VOLTAGE ERROR (%) 0.75
0
-0.2 VOLTAGE ERROR (%)
0.50
-0.4
0.25
-0.6
0
-0.8
-0.25 20 -16 -14 -12 INPUT VOLTAGE (V) -10 -8
-1.0 0 10 20 30 40 50 LOAD CURRENT (mA)
8
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TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
Typical Operating Characteristics (continued)
(Circuit of Figure 1. VIN = 5V, VMAIN = 13V, VGON = 24V, VGOFF = -8V, VOUT1 = VOUT2 = VOUT3 = VOUT4 = VOUT5 = 6.5V, TA = +25C unless otherwise noted.)
MAX1518 OPERATIONAL-AMPLIFIER SUPPLY CURRENT vs. SUPPLY VOLTAGE
MAX1516 toc13
POWER-UP SEQUENCE
MAX1516 toc12
OPERATIONAL-AMPLIFIER RAIL-TO-RAIL INPUT/OUTPUT
MAX1516 toc14
6 A SUPPLY CURRENT (mA) 0V B 0V 0V C 5 4 3 2 1 D 0V 4ms/div A: VMAIN, 10V/div B: VSRC, 20V/div C: VGOFF, 10V/div D: VGON, 20V/div 0 4.5 6.0 6.5 7.0 7.5 8.0 NO-LOAD BUFFER CONFIGURATION VPOS1 TO VPOS5 = VSUP / 2
VSUP = 6V
A
0V
B
0V 8.5 40s/div A: INPUT SIGNAL, 2V/div B: OUTPUT SIGNAL, 2V/div SUPPLY VOLTAGE (V)
OPERATIONAL-AMPLIFIER LOAD-TRANSIENT RESPONSE
MAX1516 toc15
OPERATIONAL-AMPLIFIER LARGE-SIGNAL STEP RESPONSE
MAX1516 toc16
OPERATIONAL-AMPLIFIER SMALL-SIGNAL STEP RESPONSE
MAX1516 toc17
VSUP = 6V 0V A A A 0V +50mA B 0 -50mA 0V 400ns/div A: OUTPUT VOLTAGE, 1V/div, AC-COUPLED B: OUTPUT CURRENT, 50mA/div 1s/div A: INPUT SIGNAL, 2V/div B: OUTPUT SIGNAL, 2V/div 400ns/div A: INPUT SIGNAL, 100mV/div B: OUTPUT SIGNAL, 100mV/div 0V 0V
B B
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TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
Pin Description
PIN 1 2 3 NAME MAX1516 SRC REF AGND MAX1517 SRC REF AGND MAX1518 SRC REF AGND FUNCTION Switch Input. Source of the internal high-voltage p-channel MOSFET. Bypass SRC to PGND with a minimum 0.1F capacitor close to the pins. Reference Bypass Terminal. Bypass REF to AGND with a minimum of 0.22F close to the pins. Analog Ground for Step-Up Regulator and Linear Regulators. Connect to power ground (PGND) underneath the IC. Power Ground. PGND is the source of the main step-up n-channel power MOSFET. Connect PGND to the input-capacitor ground terminals through a short, wide PC board trace. Connect to analog ground (AGND) underneath the IC. Operational-Amplifier 1 Output Operational-Amplifier 1 Inverting Input Operational-Amplifier 1 Noninverting Input Operational-Amplifier 2 Output for the MAX1518 and MAX1517. Not Internally Connected for the MAX1516. Operational-Amplifier 2 Inverting Input for the MAX1518 and MAX1517. Not Internally Connected for the MAX1516. Operational-Amplifier 2 Noninverting Input for the MAX1518 and MAX1517. Internally Connected for the MAX1516. Connect this pin to GND for the MAX1516. Analog Ground for Operational Amplifiers. Connect to power ground (PGND) underneath the IC. Operational-Amplifier 3 Noninverting Input for the MAX1518. Not Internally Connected for the MAX1517 and MAX1516. Operational-Amplifier 3 Output. Not Internally Connected for the MAX1517 and MAX1516. Operational-Amplifier Power Input. Positive supply rail for the operational amplifiers. Typically connected to VMAIN. Bypass SUP to BGND with a 0.1F capacitor. Operational-Amplifier 4 Noninverting Input for the MAX1518. Operational-Amplifier 3 Noninverting Input for the MAX1517. Not Internally Connected for the MAX1516. Operational-Amplifier 4 Inverting Input for the MAX1518. Operational-Amplifier 3 Inverting Input for the MAX1517. Not Internally Connected for the MAX1516. Operational-Amplifier 4 Output for the MAX1518. Operational-Amplifier 3 Output for the MAX1517. Not Internally Connected for the MAX1516. Operational-Amplifier 5 Noninverting Input for the MAX1518. Internally Connected for the MAX1517 and MAX1516. Connect this pin to GND for the MAX1517 and MAX1516. Operational-Amplifier 5 Inverting Input. Not Internally Connected for the MAX1517 and MAX1516.
4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19
PGND OUT1 NEG1 POS1 N.C. N.C. I. C. BGND N.C. N.C. SUP N.C. N.C. N.C. I. C. N.C.
PGND OUT1 NEG1 POS1 OUT2 NEG2 POS2 BGND N.C. N.C. SUP POS3 NEG3 OUT3 I. C. N.C.
PGND OUT1 NEG1 POS1 OUT2 NEG2 POS2 BGND POS3 OUT3 SUP POS4 NEG4 OUT4 POS5 NEG5
10
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TFT-LCD DC-DC Converters with Operational Amplifiers
Pin Description (continued)
PIN 20 21 22 23 NAME MAX1516 N.C. LX IN FB MAX1517 N.C. LX IN FB MAX1518 OUT5 LX IN FB FUNCTION Operational-Amplifier 5 Output. Not Internally Connected for the MAX1517 and MAX1516. N-Channel Power MOSFET Drain and Switching Node. Connect the inductor and Schottky diode to LX and minimize the trace area for lowest EMI. Supply Voltage Input. IN can range from 2.6V to 5.5V. Step-Up Regulator Feedback Input. Regulates to 1.236V (nominal). Connect a resistive voltage-divider from the output (VMAIN) to FB to analog ground (AGND). Place the divider within 5mm of FB. Step-Up Regulator Error-Amplifier Compensation Point. Connect a series RC from COMP to AGND. See the Loop Compensation section for component selection guidelines. Gate-On Linear-Regulator Feedback Input. FBP regulates to 1.25V (nominal). Connect FBP to the center of a resistive voltage-divider between the regulator output and AGND to set the gate-on linear-regulator output voltage. Place the resistive voltage-divider close to the pin. Gate-On Linear-Regulator Base Drive. Open drain of an internal n-channel MOSFET. Connect DRVP to the base of an external pnp pass transistor. See the Pass-Transistor Selection section. Gate-Off Linear-Regulator Feedback Input. FBN regulates to 250mV (nominal). Connect FBN to the center of a resistive voltage-divider between the regulator output and REF to set the gate-off linear-regulator output voltage. Place the resistive voltagedivider close to the pin. Gate-Off Linear-Regulator Base Drive. Open drain of an internal p-channel MOSFET. Connect DRVN to the base of an external npn pass transistor. See the Pass-Transistor Selection section. High-Voltage Switch Delay Input. Connect a capacitor from DEL to AGND to set the high-voltage switch startup delay. High-Voltage Switch Control Input. When CTL is high, the high-voltage switch between COM and SRC is on and the high-voltage switch between COM and DRN is off. When CTL is low, the high-voltage switch between COM and SRC is off and the high-voltage switch between COM and DRN is on. CTL is inhibited by the undervoltage lockout and when the voltage on DEL is less than 1.25V. Switch Input. Drain of the internal high-voltage back-to-back p-channel MOSFETs connected to COM. Internal High-Voltage MOSFET Switch Common Terminal. Do not allow the voltage on COM to exceed VSRC.
MAX1516/MAX1517/MAX1518
24
COMP
COMP
COMP
25
FBP
FBP
FBP
26
DRVP
DRVP
DRVP
27
FBN
FBN
FBN
28
DRVN
DRVN
DRVN
29
DEL
DEL
DEL
30
CTL
CTL
CTL
31 32
DRN COM
DRN COM
DRN COM
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11
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
Typical Operating Circuit
The MAX1518 Typical Operating Circuit (Figure 1) is a complete power-supply system for TFT LCDs. The circuit generates a +13V source-driver supply and +24V and -8V gate-driver supplies. The input voltage range for the IC is from +2.6V to +5.5V. The listed load currents in Figure 1 are available from a +4.5V to +5.5V supply. Table 1 lists some recommended components, and Table 2 lists the contact information of component suppliers.
Table 1. Component List
DESIGNATION C1 C2 D1 D2, D3 L1 Q1 Q2 DESCRIPTION 22F, 6.3V X5R ceramic capacitor (1210) TDK C3225X5R0J227M 22F, 16V X5R ceramic capacitor (1812) TDK C4532X5X1C226M 3A, 30V Schottky diode (M-flat) Toshiba CMS02 200mA, 100V, dual ultra-fast diodes (SOT23) Fairchild MMBD4148SE 3.0H, 3A inductor Sumida CDRH6D28-3R0 200mA, 40V pnp bipolar transistor (SOT23) Fairchild MMBT3906 200mA, 40V npn bipolar transistor (SOT23) Fairchild MMBT3904
Detailed Description
The MAX1516/MAX1517/MAX1518 contain a highperformance step-up switching regulator, two low-cost linear-regulator controllers, multiple high-current operational amplifiers, and startup timing and level-shifting functionality useful for active-matrix TFT LCDs. Figure 2 shows the MAX1518 Functional Diagram.
Main Step-Up Regulator
The main step-up regulator employs a current-mode, fixed-frequency PWM architecture to maximize loop bandwidth and provide fast transient response to pulsed loads typical of TFT-LCD panel source drivers. The 1.2MHz switching frequency allows the use of lowprofile inductors and ceramic capacitors to minimize the thickness of LCD panel designs. The integrated high-efficiency MOSFET and the IC's built-in digital soft-start functions reduce the number of external components required while controlling inrush currents. The output voltage can be set from VIN to 13V with an external resistive voltage-divider. To generate an output voltage greater than 13V, an external cascoded MOSFET is needed. See the Generating Output Voltages > 13V section in the Design Procedures. The regulator controls the output voltage and the power delivered to the output by modulating the duty cycle (D) of the internal power MOSFET in each switching cycle. The duty cycle of the MOSFET is approximated by: V -V D MAIN IN VMAIN
Figure 3 shows the Functional Diagram of the step-up regulator. An error amplifier compares the signal at FB to 1.236V and changes the COMP output. The voltage at COMP sets the peak inductor current. As the load varies, the error amplifier sources or sinks current to the COMP output accordingly to produce the inductor peak current necessary to service the load. To maintain stability at high duty cycles, a slope-compensation signal is summed with the current-sense signal. On the rising edge of the internal clock, the controller sets a flip-flop, turning on the n-channel MOSFET and applying the input voltage across the inductor. The current through the inductor ramps up linearly, storing energy in its magnetic field. Once the sum of the current-feedback signal and the slope compensation exceeds the COMP voltage, the controller resets the flip-flop and turns off the MOSFET. Since the inductor current is continuous, a transverse potential develops across the inductor that turns on the diode (D1). The voltage across the inductor then becomes the difference between the output voltage and the input voltage.
Table 2. Component Suppliers
SUPPLIER Fairchild Sumida TDK Toshiba PHONE 408-822-2000 847-545-6700 847-803-6100 949-455-2000 FAX 408-822-2102 847-545-6720 847-390-4405 949-859-3963 www.sumida.com www.component.tdk.com www.toshiba.com/taec WEBSITE www.fairchildsemi.com
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TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
VIN 4.5V TO 5.5V
L1 3.0H C1 22F R10 10 IN C18 0.1F LX FB
LX D1 C2 22F R1 95.3k 1% R1 10.2k 1% AGND LX 0.1F VMAIN 13V/500mA
180k COMP 220F LX 0.1F 6.8k 0.1F Q2 DRVN
PGND
0.1F
D2
MAX1518
DRVP
6.8k Q1
D3
VGOFF -8V/50mA 0.22F
R7 332k 1%
R4 192k 1% R5 10.0k 1%
FBP FBN 0.47F
R8 40.2k 1% REF 0.22F
SRC COM DRN DEL CTL SUP 0.1F BGND NEG1 OUT1 NEG2 OUT2 OUT3 POS3 NEG4 POS4 OUT4 POS5 NEG5 OUT5 VGON 24V/20mA
0.033F
POS1 POS2
TO VCOM BACKPLANE
Figure 1. Typical Operating Circuit ______________________________________________________________________________________ 13
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
VCN VCP
VIN LX IN STEP-UP CONTROLLER PGND COMP AGND FB
VMAIN
VCP
MAX1518
DRVP GATE-ON CONTROLLER FBP VGON
SRC DEL COM SWITCH CONTROL CTL VCN DRN
DRVN GATE-OFF CONTROLLER SUP NEG1 FBN VGOFF
OUT1
OP1 REF REF NEG4
POS1 NEG2
OUT2
OP2
OP4
OUT4
POS2
POS4 NEG5
OUT3
OP3
OP5
OUT5
POS3 BGND
POS5
Figure 2. MAX1518 Functional Diagram 14 ______________________________________________________________________________________
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
LX RESET DOMINANT CLOCK S R ILIM COMPARATOR Q PGND
SOFTSTART
VLIMIT
SLOPE COMP PWM COMPARATOR
CURRENT SENSE
OSCILLATOR FAULT COMPARATOR TO FAULT LATCH 1.0V ERROR AMP FB
1.236V COMP
Figure 3. Step-Up Regulator Functional Diagram
This discharge condition forces the current through the inductor to ramp back down, transferring the energy stored in the magnetic field to the output capacitor and the load. The MOSFET remains off for the rest of the clock cycle.
Gate-On Linear-Regulator Controller, REG P
The gate-on linear-regulator controller (REG P) is an analog gain block with an open-drain n-channel output. It drives an external pnp pass transistor with a 6.8k base-to-emitter resistor (Figure 1). Its guaranteed basedrive sink current is at least 1mA. The regulator including Q1 in Figure 1 uses a 0.47F ceramic output capacitor and is designed to deliver 20mA at 24V. Other output voltages and currents are possible with the proper pass transistor and output capacitor. See the Pass-Transistor Selection and Stability Requirements sections.
REG P is typically used to provide the TFT-LCD gate drivers' gate-on voltage. Use a charge pump with as many stages as necessary to obtain a voltage exceeding the required gate-on voltage (see the Selecting the Number of Charge-Pump Stages section). Note the voltage rating of the DRVP is 28V. If the charge-pump output voltage can exceed 28V, an external cascode npn transistor should be added as shown in Figure 4. Alternately, the linear regulator can control an intermediate charge-pump stage while regulating the final charge-pump output (Figure 5). REG P is enabled after the REF voltage exceeds 1.0V. Each time it is enabled, the controller goes through a soft-start routine that ramps up its internal reference DAC in 128 steps.
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TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
LX
VMAIN
FROM CHARGE-PUMP OUTPUT
0.1F VMAIN 13V 0.1F
DRVP NPN CASCODE TRANSISTOR
PNP PASS TRANSISTOR
6.8k
MAX1516 MAX1517 MAX1518
FBP
VGON
DRVP
Q1
VGON 35V
MAX1516 MAX1517 MAX1518
FBP
0.47F 267k 1%
0.22F
10.0k 1%
Figure 4. Using Cascoded npn for Charge-Pump Output Voltages >28V
Figure 5. The linear regulator controls the intermediate chargepump stage.
Gate-Off Linear-Regulator Controller, REG N
The gate-off linear-regulator controller (REG N) is an analog gain block with an open-drain p-channel output. It drives an external npn pass transistor with a 6.8k base-to-emitter resistor (Figure 1). Its guaranteed basedrive source current is at least 1mA. The regulator including Q2 in Figure 1 uses a 0.47F ceramic output capacitor and is designed to deliver 50mA at -8V. Other output voltages and currents are possible with the proper pass transistor and output capacitor (see the PassTransistor Selection and Stability Requirements sections). REG N is typically used to provide the TFT-LCD gate drivers' gate-off voltage. A negative voltage can be produced using a charge-pump circuit as shown in Figure 1. REG N is enabled after the voltage on REF exceeds 1.0V. Each time it is enabled, the control goes through a soft-start routine that ramps down its internal reference DAC from VREF to 250mV in 128 steps.
Short-Circuit Current Limit The operational amplifiers limit short-circuit current to approximately 150mA if the output is directly shorted to SUP or to BGND. If the short-circuit condition persists, the junction temperature of the IC rises until it reaches the thermal-shutdown threshold (+160C typ). Once the junction temperature reaches the thermal-shutdown threshold, an internal thermal sensor immediately sets the thermal fault latch, shutting off all the IC's outputs. The device remains inactive until the input voltage is cycled. Driving Pure Capacitive Load The operational amplifiers are typically used to drive the LCD backplane (VCOM) or the gamma-correction divider string. The LCD backplane consists of a distributed series capacitance and resistance, a load that can be easily driven by the operational amplifier. However, if the operational amplifier is used in an application with a pure capacitive load, steps must be taken to ensure stable operation. As the operational amplifier's capacitive load increases, the amplifier's bandwidth decreases and gain peaking increases. A 5 to 50 small resistor placed between OUT_ and the capacitive load reduces peaking but also reduces the gain. An alternative method of reducing peaking is to place a series RC network (snubber) in parallel with the capacitive load. The RC network does not continuously load the output or reduce the gain. Typical values of the resistor are between 100 and 200, and the typical value of the capacitor is 10nF.
Operational Amplifiers
The MAX1518 has five operational amplifiers, the MAX1517 has three operational amplifiers, and the MAX1516 has one operational amplifier. The operational amplifiers are typically used to drive the LCD backplane (VCOM) or the gamma-correction divider string. They feature 150mA output short-circuit current, 13V/s slew rate, and 12MHz bandwidth. The rail-to-rail input and output capability maximizes system flexibility.
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TFT-LCD DC-DC Converters with Operational Amplifiers
Undervoltage Lockout (UVLO)
The undervoltage-lockout (UVLO) circuit compares the input voltage at IN with the UVLO threshold (2.5V rising, 2.35V falling, typ) to ensure the input voltage is high enough for reliable operation. The 150mV (typ) hysteresis prevents supply transients from causing a restart. Once the input voltage exceeds the UVLO rising threshold, startup begins. When the input voltage falls below the UVLO falling threshold, the controller turns off the main step-up regulator, turns off the linear-regulator outputs, and disables the switch control block; the operationalamplifier outputs are high impedance.
VIN 2.5V 1.05V VREF
MAX1516/MAX1517/MAX1518
VMAIN
VGON
Reference Voltage (REF)
The reference output is nominally 1.25V and can source at least 50A (see the Typical Operating Characteristics). Bypass REF with a 0.22F ceramic capacitor connected between REF and AGND.
12ms INPUT SOFT- SOFTVOLTAGE START START OK BEGINS ENDS 1.25V SWITCH CONTROL ENABLED VGOFF VDEL
Power-Up Sequence and Soft-Start
Once the voltage on IN exceeds approximately 1.7V, the reference turns on. With a 0.22F REF bypass capacitor, the reference reaches its regulation voltage of 1.25V in approximately 1ms. When the reference voltage exceeds 1.0V, the ICs enable the main step-up regulator, the gate-on linear-regulator controller, and the gate-off linear-regulator controller simultaneously. The IC employs soft-start for each regulator to minimize inrush current and voltage overshoot and to ensure a well-defined startup behavior. During the soft-start, the main step-up regulator directly limits the peak inductor current. The current-limit level is increased through the soft-start period from 0 up to the full current-limit value in eight equal current steps (ILIM / 8). The maximum load current is available after the output voltage reaches regulation (which terminates soft-start), or after the soft-start timer expires. Both linear-regulator controllers use a 7-bit soft-start DAC. For the gate-on linear regulator, the DAC output is stepped in 128 steps from zero up to the reference voltage. For the gate-off linear regulator, the DAC output steps from the reference down to 250mV in 128 steps. The soft-start duration is 14ms (typ) for all three regulators. A capacitor (CDEL) from DEL to AGND determines the switch-control-block startup delay. After the input voltage exceeds the UVLO threshold (2.5V typ) and the soft-start routine for each regulator is complete and there is no fault detected, a 5A current source starts charging CDEL. Once the capacitor voltage exceeds
Figure 6. Power-Up Sequence
1.25V (typ), the switch-control block is enabled as shown in Figure 6. After the switch-control block is enabled, COM can be connected to SRC or DRN through the internal p-channel switches, depending upon the state of CTL. Before startup and when IN is less than VUVLO, DEL is internally connected to AGND to discharge CDEL. Select CDEL to set the delay time using the following equation: 5A 1.25V
CDEL = DELAY _ TIME x
Switch-Control Block
The switch-control input (CTL) is not activated until all four of the following conditions are satisfied: the input voltage exceeds VUVLO, the soft-start routine of all the regulators is complete, there is no fault condition detected, and VDEL exceeds its turn-on threshold. As shown in Figure 7, COM is pulled down to PGND through a 1k resistor when the switch control is not activated. Once activated and if CTL is high, the 5 internal p-channel switch (Q1) between COM and SRC turns on and the 30 p-channel switch (Q2) between DRN and COM turns off. If CTL is low, Q1 turns off and Q2 turns on.
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TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
IN
5A
MAX1516 MAX1517 MAX1518
2.5V
FB OK FBP OK FBN OK
Q1 DEL
SRC
REF COM
CTL
1k
Q2
Q3
DRN
Figure 7. Switch-Control Block
Fault Protection
During steady-state operation, if the output of the main regulator or any of the linear-regulator outputs does not exceed its respective fault-detection threshold, the MAX1516/MAX1517/MAX1518 activate an internal fault timer. If any condition or combination of conditions indicates a continuous fault for the fault-timer duration (55ms typ), the MAX1516/MAX1517/MAX1518 set the fault latch to shut down all the outputs except the reference. Once the fault condition is removed, cycle the input voltage (below the UVLO falling threshold) to clear the fault latch and reactivate the device. The faultdetection circuit is disabled during the soft-start time.
Thermal-Overload Protection
Thermal-overload protection prevents excessive power dissipation from overheating the MAX1516/MAX1517/ MAX1518. When the junction temperature exceeds TJ = +160C, a thermal sensor immediately activates the fault protection, which shuts down all outputs except the reference, allowing the device to cool down. Once the device cools down by approximately 15C, cycle the input voltage (below the UVLO falling threshold) to clear the fault latch and reactivate the device. The thermal-overload protection protects the controller in the event of fault conditions. For continuous operation, do not exceed the absolute maximum junction temperature rating of TJ = +150C.
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TFT-LCD DC-DC Converters with Operational Amplifiers
Design Procedure
Main Step-Up Regulator
Inductor Selection The minimum inductance value, peak current rating, and series resistance are factors to consider when selecting the inductor. These factors influence the converter's efficiency, maximum output load capability, transient-response time, and output voltage ripple. Size and cost are also important factors to consider. The maximum output current, input voltage, output voltage, and switching frequency determine the inductor value. Very high inductance values minimize the current ripple and therefore reduce the peak current, which decreases core losses in the inductor and I2R(R) losses in the entire power path. However, large inductor values also require more energy storage and more turns of wire, which increases size and can increase I 2 R losses in the inductor. Low inductance values decrease the size but increase the current ripple and peak current. Finding the best inductor involves choosing the best compromise between circuit efficiency, inductor size, and cost. The equations used here include a constant LIR, which is the ratio of the inductor peak-to-peak ripple current to the average DC inductor current at the full load current. The best trade-off between inductor size and circuit efficiency for step-up regulators generally has an LIR between 0.3 and 0.5. However, depending on the AC characteristics of the inductor core material and ratio of inductor resistance to other power-path resistances, the best LIR can shift up or down. If the inductor resistance is relatively high, more ripple can be accepted to reduce the number of turns required and increase the wire diameter. If the inductor resistance is relatively low, increasing inductance to lower the peak current can decrease losses throughout the power path. If extremely thin high-resistance inductors are used, as is common for LCD-panel applications, the best LIR can increase to between 0.5 and 1.0. Once a physical inductor is chosen, higher and lower values of the inductor should be evaluated for efficiency improvements in typical operating regions. Calculate the approximate inductor value using the typical input voltage (VIN), the maximum output current (IMAIN(MAX)), the expected efficiency (TYP) taken from an appropriate curve in the Typical Operating Characteristics section, and an estimate of LIR based on the above discussion:
I2R is a registered trademark of Instruments for Research and Industry, Inc. ______________________________________________________________________________________ 19
MAX1516/MAX1517/MAX1518
V 2 VMAIN - VIN TYP L = IN VMAIN IMAIN(MAX) x fOSC LIR Choose an available inductor value from an appropriate inductor family. Calculate the maximum DC input current at the minimum input voltage (VIN(MIN)) using conservation of energy and the expected efficiency at that operating point (MIN) taken from the appropriate curve in the Typical Operating Characteristics: IIN(DCMAX) = , IMAIN(MAX) x VMAIN VIN(MIN) x MIN
Calculate the ripple current at that operating point and the peak current required for the inductor: VIN(MIN) x (VMAIN - VIN(MIN) )
IRIPPLE =
L x VMAIN x fOSC I IPEAK = IIN(DCMAX) + RIPPLE , 2 The inductor's saturation current rating and the MAX1516/MAX1517/MAX1518s' LX current limit (ILIM) should exceed IPEAK, and the inductor's DC current rating should exceed IIN(DC,MAX). For good efficiency, choose an inductor with less than 0.1 series resistance. Considering the Typical Operating Circuit, the maximum load current (IMAIN(MAX)) is 500mA with a 13V output and a typical input voltage of 5V. Choosing an LIR of 0.5 and estimating efficiency of 85% at this operating point: 5V 2 13V - 5V 0.85 L= 3.3H 13V 0.5A x 1.2MHz 0.5 Using the circuit's minimum input voltage (4.5V) and estimating efficiency of 80% at that operating point: IIN(DCMAX) = , 0.5A x 13V 1.8A 4.5V x 0.8
The ripple current and the peak current are: IRIPPLE = 4.5V x (13V - 4.5V) 0.74A 3.3H x 13V x 1.2MHz 0.74A IPEAK = 1.8A + 2.2A 2
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
Output-Capacitor Selection The total output voltage ripple has two components: the capacitive ripple caused by the charging and discharging of the output capacitance, and the ohmic ripple due to the capacitor's equivalent series resistance (ESR). VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) V I -V VRIPPLE(C) MAIN MAIN IN , and COUT VMAINfOSC VRIPPLE(ESR) IPEAKRESR(COUT) where I PEAK is the peak inductor current (see the Inductor Selection section). For ceramic capacitors, the output voltage ripple is typically dominated by VRIPPLE(C). The voltage rating and temperature characteristics of the output capacitor must also be considered. Input-Capacitor Selection The input capacitor (CIN) reduces the current peaks drawn from the input supply and reduces noise injection into the IC. A 22F ceramic capacitor is used in the Typical Applications Circuit (Figure 1) because of the high source impedance seen in typical lab setups. Actual applications usually have much lower source impedance since the step-up regulator often runs directly from the output of another regulated supply. Typically, CIN can be reduced below the values used in the Typical Applications Circuit. Ensure a low-noise supply at IN by using adequate C IN . Alternately, greater voltage variation can be tolerated on CIN if IN is decoupled from CIN using an RC lowpass filter (see R10 and C18 in Figure 1). Rectifier Diode The MAX1516/MAX1517/MAX1518s' high switching frequency demands a high-speed rectifier. Schottky diodes are recommended for most applications because of their fast recovery time and low forward voltage. In general, a 2A Schottky diode complements the internal MOSFET well. Output-Voltage Selection The output voltage of the main step-up regulator can be adjusted by connecting a resistive voltage-divider from the output (VMAIN) to AGND with the center tap connected to FB (see Figure 1). Select R2 in the 10k to 50k range. Calculate R1 with the following equation: V R1 = R2 x MAIN - 1 VFB where VFB, the step-up regulator's feedback set point, is 1.236V. Place R1 and R2 close to the IC. Generating Output Voltages >13V The maximum output voltage of the step-up regulator is 13V, which is limited by the absolute maximum rating of the internal power MOSFET. To achieve higher output voltages, an external n-channel MOSFET can be cascoded with the internal FET (Figure 8). Since the gate of the external FET is biased from the input supply, use a logiclevel FET to ensure that the FET is fully enhanced at the minimum input voltage. The current rating of the FET needs to be higher than the IC's internal current limit. Loop Compensation Choose RCOMP to set the high-frequency integrator gain for fast transient response. Choose CCOMP to set the integrator zero to maintain loop stability. For low-ESR output capacitors, use the following equations to obtain stable performance and good transient response: RCOMP CCOMP 315 x VIN x VOUT x COUT L x IMAIN(MAX) VOUT x COUT 10 x IMAIN(MAX) x RCOMP
To further optimize transient response, vary RCOMP in 20% steps and CCOMP in 50% steps while observing transient-response waveforms.
Charge Pumps
Selecting the Number of Charge-Pump Stages For highest efficiency, always choose the lowest number of charge-pump stages that meet the output requirement. Figures 9 and 10 show the positive and negative charge-pump output voltages for a given VMAIN for one-, two-, and three-stage charge pumps. The number of positive charge-pump stages is given by: V +V -V nPOS = GON DROPOUT MAIN VMAIN - 2 x VD where nPOS is the number of positive charge-pump stages, VGON is the gate-on linear-regulator REG P output, VMAIN is the main step-up regulator output, VD is the forward-voltage drop of the charge-pump diode, and VDROPOUT is the dropout margin for the linear regulator. Use VDROPOUT = 0.3V.
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TFT-LCD DC-DC Converters with Operational Amplifiers
VIN VMAIN >13V
LX FB STEP-UP CONTROLLER PGND
nitely has a negligible effect on output-current capability because the internal switch resistance and the diode impedance place a lower limit on the source impedance. A 0.1F ceramic capacitor works well in most low-current applications. The flying capacitor's voltage rating must exceed the following: VCX > n x VMAIN where n is the stage number in which the flying capacitor appears, and VMAIN is the output voltage of the main step-up regulator. Charge-Pump Output Capacitor Increasing the output capacitance or decreasing the ESR reduces the output ripple voltage and the peak-topeak transient voltage. With ceramic capacitors, the output voltage ripple is dominated by the capacitance value. Use the following equation to approximate the required capacitor value: COUT _ CP ILOAD _ CP 2fOSC VRIPPLE _ CP
MAX1516/MAX1517/MAX1518
MAX1516 MAX1517 MAX1518
Figure 8. Operation with Output Voltages >13V Using Cascoded MOSFET
The number of negative charge-pump stages is given by: -V + VDROPOUT nNEG = GOFF VMAIN - 2 x VD where nNEG is the number of negative charge-pump stages, VGOFF is the gate-off linear-regulator REG N output, VMAIN is the main step-up regulator output, VD is the forward-voltage drop of the charge-pump diode, and VDROPOUT is the dropout margin for the linear regulator. Use VDROPOUT = 0.3V. The above equations are derived based on the assumption that the first stage of the positive charge pump is connected to VMAIN and the first stage of the negative charge pump is connected to ground. Sometimes fractional stages are more desirable for better efficiency. This can be done by connecting the first stage to VIN or another available supply. If the first charge-pump stage is powered from V IN , then the above equations become: V +V +V nPOS = GON DROPOUT IN VMAIN - 2 x VD -V + VDROPOUT + VIN nNEG = GOFF VMAIN - 2 x VD Flying Capacitors Increasing the flying-capacitor (CX) value lowers the effective source impedance and increases the outputcurrent capability. Increasing the capacitance indefi-
where COUT_CP is the output capacitor of the charge pump, I LOAD_CP is the load current of the charge pump, and VRIPPLE_CP is the peak-to-peak value of the output ripple. Charge-Pump Rectifier Diodes Use low-cost silicon switching diodes with a current rating equal to or greater than two times the average charge-pump input current. If it helps avoid an extra stage, some or all of the diodes can be replaced with Schottky diodes with an equivalent current rating.
Linear-Regulator Controllers
Output-Voltage Selection Adjust the gate-on linear-regulator (REG P) output voltage by connecting a resistive voltage-divider from the REG P output to AGND with the center tap connected to FBP (Figure 1). Select the lower resistor of the divider R5 in the range of 10k to 30k. Calculate the upper resistor R4 with the following equation: V R4 = R5 x GON - 1 VFBP where VFBP = 1.25V (typ).
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TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
POSITIVE CHARGE-PUMP OUTPUT VOLTAGE vs. VMAIN
60 VD = 0.3V TO 1V 50 40 2-STAGE CHARGE PUMP 30 20 10 1-STAGE CHARGE PUMP 0 2 4 6 8 VMAIN (V) 10 12 14 G_OFF (V) G_ON (V) 3-STAGE CHARGE PUMP -0 -5 -10 -15 -20 -25 -30 -35 -40 -45 2 4 6 8 VMAIN (V) 10 12 14 2-STAGE CHARGE PUMP 3-STAGE CHARGE PUMP VD = 0.3V TO 1V 1-STAGE CHARGE PUMP
NEGATIVE CHARGE-PUMP OUTPUT VOLTAGE vs. VMAIN
Figure 9. Positive Charge-Pump Output Voltage vs. VMAIN
Figure 10. Negative Charge-Pump Output Voltage vs. VMAIN
Adjust the gate-off linear-regulator REG N output voltage by connecting a resistive voltage-divider from VGOFF to REF with the center tap connected to FBN (Figure 1). Select R8 in the range of 20k to 50k. Calculate R7 with the following equation: V -V R7 = R8 x FBN GOFF VREF - VFBN where VFBN = 250mV, VREF = 1.25V. Note that REF can only source up to 50A; using a resistor less than 20k for R8 results in higher bias current than REF can supply. Pass-Transistor Selection The pass transistor must meet specifications for current gain (hFE), input capacitance, collector-emitter saturation voltage and power dissipation. The transistor's current gain limits the guaranteed maximum output current to: V ILOAD(MAX) = IDRV - BE x hFE(MIN) RBE where IDRV is the minimum guaranteed base-drive current, VBE is the transistor's base-to-emitter forward voltage drop, and RBE is the pullup resistor connected between the transistor's base and emitter. Furthermore, the transistor's current gain increases the linear regulator's DC loop gain (see the Stability Requirements section), so excessive gain destabilizes the output.
Therefore, transistors with current gain over 100 at the maximum output current can be difficult to stabilize and are not recommended unless the high gain is needed to meet the load-current requirements. The transistor's saturation voltage at the maximum output current determines the minimum input-to-output voltage differential that the linear regulator can support. Also, the package's power dissipation limits the usable maximum input-to-output voltage differential. The maximum power-dissipation capability of the transistor's package and mounting must exceed the actual power dissipated in the device. The power dissipated equals the maximum load current (ILOAD(MAX)_LR) multiplied by the maximum input-to-output voltage differential: P = ILOAD(MAX)_ LR x (VIN(MAX)_ LR - VOUT _ LR ) where VIN(MAX)_LR is the maximum input voltage of the linear regulator, and VOUT_LR is the output voltage of the linear regulator. Stability Requirements The MAX1516/MAX1517/MAX1518 linear-regulator controllers use an internal transconductance amplifier to drive an external pass transistor. The transconductance amplifier, the pass transistor, the base-emitter resistor, and the output capacitor determine the loop stability. The following applies to both linear-regulator controllers in the MAX1516/MAX1517/MAX1518.
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TFT-LCD DC-DC Converters with Operational Amplifiers
The transconductance amplifier regulates the output voltage by controlling the pass transistor's base current. The total DC loop gain is approximately: 10 I xh A V _ LR x 1 + BIAS FE x VREF VT ILOAD _ LR where VT is 26mV at room temperature, and IBIAS is the current through the base-to-emitter resistor (RBE). For the MAX1516/MAX1517/MAX1518, the bias currents for both the gate-on and gate-off linear-regulator controllers are 0.1mA. Therefore, the base-to-emitter resistor for both linear regulators should be chosen to set 0.1mA bias current: RBE = 0.7V VBE = 6.8k 0.1mA 0.1mA gm is the transconductance of the pass transistor, and fT is the transition frequency. Both parameters can be found in the transistor's data sheet. Because RBE is much greater than RIN, the above equation can be simplified: fPOLE _ IN = 1 2 x CIN x RIN
MAX1516/MAX1517/MAX1518
Substituting for CIN and RIN yields: f fPOLE _ IN = T hFE 4) Next, calculate the pole set by the linear regulator's feedback resistance and the capacitance between FB_ and AGND (including stray capacitance): fPOLE _ FB = 1 2 x CFB x (RUPPER || RLOWER )
The output capacitor and the load resistance create the dominant pole in the system. However, the internal amplifier delay, pass transistor's input capacitance, and the stray capacitance at the feedback node create additional poles in the system, and the output capacitor's ESR generates a zero. For proper operation, use the following equations to verify the linear regulator is properly compensated: 1) First, determine the dominant pole set by the linear regulator's output capacitor and the load resistor: fPOLE _ LR = ILOAD(MAX)_ LR 2 x COUT _ LR x VOUT _ LR
where C FB is the capacitance between FB_ and AGND, RUPPER is the upper resistor of the linear regulator's feedback divider, and RLOWER is the lower resistor of the divider. 5) Next, calculate the zero caused by the output capacitor's ESR: fPOLE _ ESR = 1 2 x COUT _ LR x RESR
The unity-gain crossover of the linear regulator is: fCROSSOVER = AV_LR fPOLE_LR 2) The pole created by the internal amplifier delay is approximately 1MHz: fPOLE_AMP = 1MHz 3) Next, calculate the pole set by the transistor's input capacitance, the transistor's input resistance, and the base-to-emitter pullup resistor: fPOLE _ IN = 1 2 x CIN x (RBE || RIN ) where RESR is the equivalent series resistance of COUT_LR. To ensure stability, choose COUT_LR large enough so the crossover occurs well before the poles and zero calculated in steps 2 to 5. The poles in steps 3 and 4 generally occur at several megahertz, and using ceramic capacitors ensures the ESR zero occurs at several megahertz as well. Placing the crossover below 500kHz is sufficient to avoid the amplifier-delay pole and generally works well, unless unusual component choices or extra capacitances move one of the other poles or the zero below 1MHz.
where CIN =
gm h , RIN = FE , 2fT gm
______________________________________________________________________________________
23
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
Applications Information
Power Dissipation
An IC's maximum power dissipation depends on the thermal resistance from the die to the ambient environment and the ambient temperature. The thermal resistance depends on the IC package, PC board copper area, other thermal mass, and airflow. The MAX1516/MAX1517/MAX1518, with their exposed backside pad soldered to 1in2 of PC board copper, can dissipate about 1.7W into +70C still air. More PC board copper, cooler ambient air, and more airflow increase the possible dissipation, while less copper or warmer air decreases the IC's dissipation capability. The major components of power dissipation are the power dissipated in the step-up regulator and the power dissipated by the operational amplifiers. Step-Up Regulator The largest portions of power dissipation in the step-up regulator are the internal MOSFET, the inductor, and the output diode. If the step-up regulator has 90% efficiency, about 3% to 5% of the power is lost in the internal MOSFET, about 3% to 4% in the inductor, and about 1% in the output diode. The remaining 1% to 3% is distributed among the input and output capacitors and the PC board traces. If the input power is about 5W, the power lost in the internal MOSFET is about 150mW to 250mW. Operational Amplifier The power dissipated in the operational amplifiers depends on their output current, the output voltage, and the supply voltage: PDSOURCE = IOUT _(SOURCE) x (VSUP - VOUT _ ) PDSINK = IOUT _(SINK) x VOUT _ where IOUT_(SOURCE) is the output current sourced by the operational amplifier, and IOUT_(SINK) is the output current that the operational amplifier sinks. In a typical case where the supply voltage is 13V and the output voltage is 6V with an output source current of 30mA, the power dissipated is 180mV. to the inductor, to the IC's LX pin, out of PGND, and to the input capacitor's negative terminal. The highcurrent output loop is from the positive terminal of the input capacitor to the inductor, to the output diode (D1), and to the positive terminal of the output capacitors, reconnecting between the output capacitor and input capacitor ground terminals. Connect these loop components with short, wide connections. Avoid using vias in the high-current paths. If vias are unavoidable, use many vias in parallel to reduce resistance and inductance. * Create a power-ground island (PGND) consisting of the input and output capacitor grounds, PGND pin, and any charge-pump components. Connect all of these together with short, wide traces or a small ground plane. Maximizing the width of the powerground traces improves efficiency and reduces output voltage ripple and noise spikes. Create an analog ground plane (AGND) consisting of the AGND pin, all the feedback-divider ground connections, the operational-amplifier divider ground connections, the COMP and DEL capacitor ground connections, and the device's exposed backside pad. Connect the AGND and PGND islands by connecting the PGND pin directly to the exposed backside pad. Make no other connections between these separate ground planes. * Place all feedback voltage-divider resistors as close to their respective feedback pins as possible. The divider's center trace should be kept short. Placing the resistors far away causes their FB traces to become antennas that can pick up switching noise. Take care to avoid running any feedback trace near LX or the switching nodes in the charge pumps. * Place the IN pin and REF pin bypass capacitors as close to the device as possible. The ground connection of the IN bypass capacitor should be connected directly to the AGND pin with a wide trace. * Minimize the length and maximize the width of the traces between the output capacitors and the load for best transient responses. * Minimize the size of the LX node while keeping it wide and short. Keep the LX node away from feedback nodes (FB, FBP, and FBN) and analog ground. Use DC traces to shield if necessary. Refer to the MAX1518 evaluation kit for an example of proper PC board layout.
PC Board Layout and Grounding
Careful PC board layout is important for proper operation. Use the following guidelines for good PC board layout: * Minimize the area of high-current loops by placing the inductor, the output diode, and the output capacitors near the input capacitors and near the LX and PGND pins. The high-current input loop goes from the positive terminal of the input capacitor
24
Chip Information
TRANSISTOR COUNT: 4608 PROCESS: BiCMOS
______________________________________________________________________________________
TFT-LCD DC-DC Converters with Operational Amplifiers
Pin Configurations
MAX1516/MAX1517/MAX1518
DRVN
DRVP
COM
DRN
FBN
32 SRC REF AGND PGND OUT1 NEG1 POS1 N.C. 1 2 3 4 5 6 7 8 9 N.C.
31
30
29
DEL
CTL
28
27
26
FBP
TOP VIEW
25 24 23 22 21 COMP FB IN LX N.C. N.C. I.C. N.C.
MAX1516
20 19 18 17
10 I.C.
11
BGND
12 N.C.
13 N.C.
14
SUP
15
N.C.
16
N.C.
THIN QFN 5mm x 5mm
N.C. = NOT INTERNALLY CONNECTED I.C. = INTERNALLY CONNECTED
DRVN
DRVP
DRVN
COM
DRVP
DRN
COM
FBN
DEL
CTL
FBP
DRN
TOP VIEW
FBN
32 SRC REF AGND PGND OUT1 NEG1 POS1 OUT2 1 2 3 4 5 6 7 8 9 NEG2
31
30
29
28
27
26
25 24 23 22 21 COMP FB IN LX N.C. N.C. I.C. OUT3
32 SRC REF AGND PGND OUT1 NEG1 POS1 OUT2 1 2 3 4 5 6 7 8 9 NEG2
31
30
29
DEL
CTL
28
27
26
FBP
TOP VIEW
25 24 23 22 21 COMP FB IN LX OUT5 NEG5 POS5 OUT4
MAX1517
20 19 18 17
MAX1518
20 19 18 17
10 POS2
11
BGND
12 N.C.
13 N.C.
14
SUP
15
POS3
16
NEG3
10 POS2
11
BGND
12 POS3
13 OUT3
14
SUP
15
POS4
16
NEG4
THIN QFN 5mm x 5mm
N.C. = NOT INTERNALLY CONNECTED I.C. = INTERNALLY CONNECTED
THIN QFN 5mm x 5mm
______________________________________________________________________________________
25
TFT-LCD DC-DC Converters with Operational Amplifiers MAX1516/MAX1517/MAX1518
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
QFN THIN.EPS
L
0.15 C A
D2
C L
D
b D2/2
0.10 M C A B
PIN # 1 I.D.
D/2
0.15 C B
k
PIN # 1 I.D. 0.35x45
E/2 E2/2 E (NE-1) X e
C L
E2
k L
DETAIL A
e (ND-1) X e
DETAIL B
e
L1
L
C L
C L
L
e 0.10 C A 0.08 C
e
C
A1
A3
PACKAGE OUTLINE 16, 20, 28, 32, 40L, THIN QFN, 5x5x0.8mm
21-0140
E
1
2
COMMON DIMENSIONS PKG. 20L 5x5 28L 5x5 32L 5x5 40L 5x5 16L 5x5 SYMBOL MIN. NOM. MAX. MIN. NOM. MAX. MIN. NOM. MAX. MIN. NOM. MAX. MIN. NOM. MAX. A A1 A3 b D E e k L L1 N ND NE JEDEC 0.70 0.75 0.80 0.70 0.75 0.80 0.70 0.75 0.80 0.70 0.75 0.80 0.70 0.75 0.80 0 0.02 0.05 0.20 REF. 0 0.02 0.05 0.20 REF. 0 0.02 0.05 0.20 REF. 0 0.02 0.05 0.20 REF. 0 0.05 0.20 REF. PKG. CODES T1655-1 T1655-2 T2055-2 T2055-3 T2055-4 T2855-1 T2855-2 T2855-3 T2855-4 T2855-5 T2855-6 T2855-7 T3255-2 T3255-3 T3255-4 T4055-1
EXPOSED PAD VARIATIONS
D2
MIN. NOM. MAX. MIN.
E2 3.10 3.20 3.10 3.20 3.10 3.20 3.10 3.20 3.10 3.25 2.70 3.25 2.70 2.70 3.25 2.70 3.10 3.10 3.10 3.20 3.35 2.80 3.35 2.80 2.80 3.35 2.80 3.20 3.20 3.20
NOM. MAX. ALLOWED
DOWN BONDS
3.00 3.00 3.00 3.00 3.00 3.15 2.60 3.15 2.60 2.60 3.15 2.60 3.00 3.00 3.00 3.20
3.10 3.20 3.00 3.10 3.20 3.00 3.10 3.20 3.00 3.10 3.20 3.00 3.10 3.25 2.70 3.25 2.70 2.70 3.25 2.70 3.10 3.10 3.10 3.20 3.35 2.80 3.35 2.80 2.80 3.35 2.80 3.20 3.20 3.20 3.00 3.15 2.60 3.15 2.60 2.60 3.15 2.60 3.00 3.00 3.00
0.25 0.30 0.35 0.25 0.30 0.35 0.20 0.25 0.30 0.20 0.25 0.30 0.15 0.20 0.25 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 0.80 BSC. 0.65 BSC. 0.50 BSC. 0.50 BSC. 0.40 BSC. - 0.25 - 0.25 - 0.25 0.35 0.45 0.25 - 0.25 0.30 0.40 0.50 0.45 0.55 0.65 0.45 0.55 0.65 0.30 0.40 0.50 0.40 0.50 0.60 16 4 4 WHHB 20 5 5 WHHC 28 7 7 WHHD-1 32 8 8 WHHD-2 0.30 0.40 0.50 40 10 10 -
NO YES NO YES NO NO NO YES YES NO NO YES NO YES NO YES
3.30 3.40 3.20
3.30 3.40
NOTES: 1. DIMENSIONING & TOLERANCING CONFORM TO ASME Y14.5M-1994. 2. ALL DIMENSIONS ARE IN MILLIMETERS. ANGLES ARE IN DEGREES. 3. N IS THE TOTAL NUMBER OF TERMINALS. 4. THE TERMINAL #1 IDENTIFIER AND TERMINAL NUMBERING CONVENTION SHALL CONFORM TO JESD 95-1 SPP-012. DETAILS OF TERMINAL #1 IDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE ZONE INDICATED. THE TERMINAL #1 IDENTIFIER MAY BE EITHER A MOLD OR MARKED FEATURE. 5. DIMENSION b APPLIES TO METALLIZED TERMINAL AND IS MEASURED BETWEEN 0.25 mm AND 0.30 mm FROM TERMINAL TIP. 6. ND AND NE REFER TO THE NUMBER OF TERMINALS ON EACH D AND E SIDE RESPECTIVELY. 7. DEPOPULATION IS POSSIBLE IN A SYMMETRICAL FASHION. 8. COPLANARITY APPLIES TO THE EXPOSED HEAT SINK SLUG AS WELL AS THE TERMINALS. 9. DRAWING CONFORMS TO JEDEC MO220, EXCEPT EXPOSED PAD DIMENSION FOR T2855-1, T2855-3 AND T2855-6. 10. WARPAGE SHALL NOT EXCEED 0.10 mm. PACKAGE OUTLINE 16, 20, 28, 32, 40L, THIN QFN, 5x5x0.8mm
21-0140
E
2
2
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
26 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2004 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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